A low-cost dual-polarized microstrip antenna array

A LOW-COST DUAL-POLARIZED MICROSTRIP ANTENNA ARRAY Shun-Shi Zhong,1 Xue-Xia Yang,1 Shi-Chang Gao,1 and Mohammed Ahmed1 1 Department of Communications Engineering Shanghai University Jiading, Shanghai 201800, P.R. China Recei¨ ed 7 July 1999; re¨ ised 14 September 1999 ABSTRACT: A new low-cost dual-polarized microstrip antenna array with a single layer is proposed. Its element is the parallel corner-fed square patch with two ports. The design, analysis, and experimental results of a 12 GHz 16-element array on a 0.8 mm thick substrate are presented. Its measured isolation is 26.5 dB at 12 GHz and higher than 20 dB o¨ er 710 MHz. The 2:1 VSWR bandwidth is wider than 840 MHz (7%). The theoretical results agree well with the experimental results. 䊚 2000 John Wiley & Sons, Inc. Microwave Opt Technol Lett 24: 176᎐179, 2000. Key words: microstrip antenna array; dual polarization; low cost; radiation patterns; S-parameters 1. INTRODUCTION

Dual-polarized antennas are required in many wireless communication systems. For satellite communication systems, the use of dual polarization doubles the system capacity. In land mobile communication systems, the polarization diversity involving the use of dual-polarized antennas combats the multipath fading and improves system performances significantly. To realize small-size terminals for mobile communications, it is necessary to integrate the receiving and transmitting functions into one antenna, while the dual-polarized antenna is employed to reduce the transmit᎐receive interference. There are two ways of realizing dual-polarized microstrip antenna arrays. One way is to use two different arrays with orthogonal polarizations w1x. The other way is to use one single array of dual-polarized elements. The most common patch element is the square patch with two feed ports at central points of orthogonal edges. Both the element itself and its array often achieve an isolation of 15᎐20 dB w2x, except that connected by wire bonds w3x. In order to improve the isolation, several patch arrays using multilayer configurations have been reported recently with an isolation of more than 20᎐30 dB w4, 5x. To save space, weight, and cost, microstrip arrays with single-layer structures are sometimes desirable. In our earlier work w6x, a 2 = 2 single-layer dual-polarized microstrip array of serial corner-fed square patches was suggested, where an isolation of 40 dB was achieved. In this paper, a new compact single-layer 4 = 4 dual-polarized array of parallel corner-fed square patches w7x is proposed. The design, analysis, and experimental results of the array are presented in order. 2. DESIGN AND ANALYSIS

2.1. Array Configuration. Figure 1 shows the layout of the 16-element dual-polarized array. Four corner-fed square patches form a series linear array by using the half-wave microstrip lines with high characteristic impedance Ž80 ⍀ in Contract grant sponsor: Natural Science Foundation of China Contract grant sponsor: Department of Antennas and Servomechanism, Communications, Telemetry & Telecontrol Research Institute, Shijiazhuang, P.R. China

176

Figure 1

16-element dual-polarized array

the present design.. Then the four-element series arrays are combined by a corporate-fed network. Quarter-wave transformers are used to transform the input impedance to 50 ⍀ at the feed point HrV. The advantages of this hybrid array are the simplicity of its structure, low loss of the series feed line of the linear array, and no beam squint in the corporate feed plane. To obtain high isolation, the corner feeding is adopted here, and every element is excited at two orthogonal corners by two symmetric feed networks. To reduce the undesirable radiation from corporate-fed networks, rounded corners are used instead of vertical bends. A 16-element array is designed to resonate at 12.0 GHz, and is fabricated on a home-grown double-sided copper-clad PTFE laminate of relative dielectric constant 2.78 and thickness 0.8 mm. The size of the patch is 7.3 mm square, and the length of the microstrip line between two adjacent patches is 8.48 mm with a width of 0.8 mm. 2.2. Theory. Since the thickness of the substrate is much less than the wavelength, the cavity model can be applied to calculate far fields. As the feed point is located at the corner of a square patch, TM 01- and TM 10-modes are excited simultaneously with the same phase and amplitude. For the TM 01mode, based on the cavity model theory w8x, its far fields of a square patch can be derived as w9x

E␪ s j

4 k 0V01 eyjk 0 r e jŽ uq¨ r2.

␭r

sin

u 2

cos

¨

2

ž

a2 u2

q

a2 ¨2 y ␲2

/

= sin ␪ sin Ž ␾ y 45T . cos Ž ␾ y 45T . E␾ s j

4 k 0V01 eyjk 0 r e jŽ uq¨ r2.

␭r y

sin

u 2

cos

¨

2

ž

a2 cos 2 Ž ␾ y 45T .

a2 sin 2 Ž ␾ y 45T . ¨2 y ␲2

u2

/

sin ␪ cos ␪

Ž1.

where V01 is the voltage of TM 01 at the corner and a is the length of the square patch, with u s k 0 a sin ␪ cos Ž ␾ y 45T . , ¨ s k 0 a sin ␪ sin Ž ␾ y 45T . . Ž2. For the TM 10-mode, similar results are obtained by interchanging u and ¨ . Then, the far fields of the corner-fed square patch are achieved as the summation of those of the TM 01- and TM 10-modes, neglecting the radiation from both of the other modes in each element and the microstrip feedline current. Finally, the far fields of the 4 = 4 array for

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 24, No. 3, February 5 2000

vertical polarization are given as E␪ s j

4 k 0V01 eyjk 0 r e jŽ uq¨ .r2

␭r u

qcos

¨

sin

2

2

a2

ž

2

¨

cos

a2

q

¨2

u

sin

u2 y ␲ 2

ž

2

a2 u2

q

a2

/

¨2 y ␲2

/

=sin ␪ sin Ž ␾ y 45T . cos Ž ␾ y 45T . = A f x A f y E␾ s j

4 k 0V01 eyjk 0 r e jŽ uq¨ .r2

␭r 2Ž

a2 sin 2 Ž ␾ y 45T .

ž

2

q cos

/

¨2 y ␲2

= y

u

T

a sin ␾ y 45 . 2

y

sin

¨

2

q

¨

cos u

u2

Z p1 , p1 V3 ᎐ 6 s V7 ᎐ 10 Z p2 , p1

¨

sin

2

ž

2

a2 cos 2 Ž ␾ y 45T .

polarized array is divided into several smaller segments with regular geometries as shown in Figure 2. These segments are analyzed individually by using a combination of the cavity model, the multiport network model, and segmentation and desegmentation techniques. Then these segments are combined one by one so as to obtain the total impedance matrix. The segmentation technique is used to combine the impedance matrix after neglecting the mutual coupling and the mutual coupling network matrix, resulting in the following impedance matrix of segment ␣ : Z p1 , p2 Z p2 , p2

I3 ᎐ 6 . I7 ᎐ 10

2

a2 cos 2 Ž ␾ y 45T . u2 y ␲ 2

/

=sin ␪ cos ␪ = A f x A f y

Ž3.

with A f x s cos

ž

k 0 S1 2

sin ␪ cos ␾ q cos

/

3k 0 S1

ž

2

sin ␪ cos ␾

/ Ž4.

A f y s cos

ž

k 0 S1 2

sin ␪ sin ␾ q cos

/

ž

3k 0 S1 2

sin ␪ sin ␾

/

Ž5.

where A f x and A f y are the array factors and S1 is the distance between two adjacent patches. In the present expressions, the uniform excitation of the array is assumed, omitting the mutual couplings between elements. From the above, the E-plane Ž ␾ s 90T . and H-plane Ž ␾ s 0T . radiation patterns will be obtained. For the calculation of cross-polarization patterns, we consider the radiation of three modes: TM 11, TM 02 , and TM 20 . The pattern factors are given as follows: FHC Ž ␪ . s

1

V11 cos 2

V01

y2V02 sin u FEC Ž ␪ . s

1

ž

2

2

u y␲2 2

1 u y␲ 2

yV11 cos 2

V01

qV02 sin u

ž

u

ž /

1 u2 y ␲ 2

2

q

1

u

1

2

u y␲2

ž /

y

sin ␪ A f x

Ž6.

cos ␪ sin ␪ AXf y

Ž7.

u y 4␲ 2 2

/

2

1 u 2 y 4␲ 2

/

with AXf y s sin

ž

k 0 S1 2

sin ␪ q sin

/

ž

3k 0 S1 2

sin ␪

/

Ž8.

where FHC Ž ␪ . and FEC Ž ␪ . are the cross-polarization patterns in the H-plane and E-plane, respectively. Vmn is the voltage of TM mn at the corner. The radiation patterns for horizontal polarization can be obtained in the same way. In order to obtain the S-parameters of the dual-polarized array, a theoretical method, called the extended multiport network ŽEMN. method, is proposed w10x. First, the dual-

Figure 2

Division of the array

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177

Where V3 , V4 , V5 , V6 and I3 , I4 , I5 , I6 are voltages and currents of ports 3᎐6, respectively, and V7 , V8 , V9 , V10 and I7 , I8 , I9 , I10 are voltages and currents of ports 7᎐10, respectively. Segments ␤ and ␥ are further divided into simpler segments with known Green’s functions. By using the multiport connection technique, their impedance matrices are derived as V1 V11 ᎐ 14 V2 V15 ᎐ 18

s

s

Z1, 1

Z1, p3

Z p3 , 1

Z p3 , p3

I1 I11 ᎐ 14

Z2, 2

Z2, p4

I2

Z p4 , 2

Z p4 , p4

I15 ᎐ 18

.

By using the continuity of tangential electric fields and magnetic fields, segments ␣ and ␤ are combined, resulting in Z1,␣q1 ␤ V1 s V7 ᎐ 10 Z 7␣᎐q10␤, 1

Z1,␣ q7 ᎐␤10

I1

Z 7␣᎐q10␤, 7 ᎐ 10

I7 ᎐ 10

.

This result is further combined with segment ␥ ; then the Z-matrix of the whole array is found as follows: array Z1, V1 1 s array V2 Z2, 1

array Z1, 2 array Z2, 2

I1 I2

Figure 3 E-plane patterns for vertical polarization. ᎐ ᎐ copolarizedrmeas. ᎐⭈᎐ cross-polarizationrmeas. ᎏᎏ copolarizedrcalc. - - - cross-polarizedrcalc.

Ž9.

where array x w Z1, s w Z1,␣q1 ␤ x y w Z1,␣q7 ᎐␤10 x 1

= Z 7␣᎐q10␤, 7 ᎐ 10 q Z p4 , p4 array x w Z1, s w Z1,␣ q7 ᎐␤10 x Z 7␣᎐q10␤, 7 ᎐ 10 q Z p4 , p4 2 array x w Z2, s w Z2, p4 x Z 7␣᎐q10␤, 7 ᎐ 10 q Z p4 , p4 1

y1

y1

y1

w Z 7␣᎐q10␤, 1 x

w Z p4 , 2 x

w Z 7␣᎐q10␤, 1 x

array x w Z2, s w Z2, 2 x y w Z2, p4 x 2

= Z 7␣᎐q10␤, 7 ᎐ 10 q Z p4 , p4

y1

w Z p4 , 2 x .

Finally, the S-parameters of the array are obtained by transforming the impedance matrix into the scattering matrix. 3. ARRAY PERFORMANCES

The E-plane and H-plane radiation patterns at 12.0 GHz for vertical polarization are shown in Figures 3 and 4, respectively. It is seen that the measured and calculated results coincide well for both co- and cross polarization. For vertical polarization in the E-plane, the cross polarization level at boresight is measured to be y23 dB down from the copolarization level, with the worst cross-polarization level of y16 dB. In the H-plane the cross-polarization level at boresight is measured to be y29 dB, with the worst cross-polarization level of y14.3 dB. In general, the cross-polarization level is lower inside the range of the main lobe. Due to the symmetry at two input ports, the patterns for both polarizations are quite alike. For horizontal polarization, the measured cross-polarization levels at boresight are y24 and y28 dB down from the copolarization level in the E-plane and H-plane, respectively. The measured worst cross-polarization level is about y15.0 dB. The measured and calculated results of VSWR at two input ports are given in Figure 5. A relatively wider band-

178

Figure 4 H-plane patterns for vertical polarization. ᎐ ᎐ copolarizedrmeas. ᎐⭈᎐ cross-polarizationrmeas. ᎏᎏ copolarizedrcalc. - - - cross-polarizedrcalc.

width of VSWR around a center frequency of 12.0 GHz is observed. The 2:1 VSWR bandwidths are measured to be 840 MHz Ž7.0%. and 890 MHz Ž7.4%. for the H- and V-ports, respectively. These are good for a single-layer microstrip antenna array of 0.8 mm thickness. Owing to the symmetry of the array at both input ports, the calculated VSWRs at the two ports are the same. A general agreement between the measured and calculated bandwidths is observed, which validates the theory. Figure 6 shows the measured and calculated results of isolation between two input ports, demonstrating good agreement between both results. The measured isolation is 26.5 dB at 12.0 GHz and higher than 20 dB over 710 MHz Žfrom 11.64 to 12.35 GHz.. The results are substantially better than those of the dual-polarized arrays with edge-fed square patches w2x.

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 24, No. 3, February 5 2000

Figure 5

VSWR at two input ports

REFERENCES 1. P. Brachat and J.M. Baracco, Dual-polarization slot-coupled printed antennas fed by stripline, IEEE Trans Antennas Propagat 43 Ž1995., 738᎐742. 2. A.G. Derneryd, Microstrip Array Antenna, Proc 6th European Microwave Conference, 1976, 339᎐343. 3. M.J. Cryan and P.S. Hall, Integrated active antenna with simultaneous transmit-receive operation, Electron Lett 32 Ž1996., 286᎐ 287. 4. Y. Murakami, J. Chiba, and Y. Karasawa, Slot-coupled self-diplexing array antenna for mobile satellite communications, Proc Inst Elect Eng 143 Ž1996., 119᎐123. 5. B. Lindmark et al., Dual-polarized aperture-coupled patch multibeam antenna, IEEE AP-S Int Symp Dig, Atlanta, GA, 1998, pp. 328᎐331. 6. S.C. Gao and S.S. Zhong, Dual-polarized microstrip antenna array with high isolation fed by coplanar network, Microwave Opt Technol Lett 19 Ž1998., 214᎐216. 7. E.M. Cruz and J.P. Daniel, Experimental analysis of corner-fed printed square patch antennas, Electron Lett 27 Ž1991., 1410᎐ 1412. 8. W.F. Richards, Y.T. Lo, and D.D. Harrison, An improved theory for microstrip antennas and applications, IEEE Trans Antennas Propagat AP-29 Ž1981., 38᎐47. 9. S.S. Zhong, Microstrip antenna theory and applications, Xidian University Press, P.R. China, 1991. 10. S.C. Gao, Microstrip antenna elements and dual-polarized arrays for active integration, Ph.D. dissertation, Shanghai University, P.R. China, 1999. 䊚 2000 John Wiley & Sons, Inc. CCC 0895-2477r00

PARALLEL EFFICIENT THREE-DIMENSIONAL BEAM PROPAGATION METHOD USING THE DU FORT – FRANKEL TECHNIQUE Husain M. Masoudi1 and John M. Arnold 2 Department of Electrical Engineering King Fahd University of Petroleum and Minerals Dhahran, Saudi Arabia 2 Department of Electronics and Electrical Engineering University of Glasgow Glasgow, Scotland

1

Figure 6

Isolation between two iput ports

Recei¨ ed 24 August 1999 4. CONCLUSION

A new low-cost dual-polarized array of corner-fed patches is presented. Its advantages are small size, low cost, higher isolation, lower cross polarization, etc. The radiation patterns for both co- and cross polarizations are formulated and calculated. Its S-parameters are found by using the CADoriented EMN method for thin microstrip antennas, which combines the cavity model, the multiport network model, and segmentation and desegmentation techniques. The theoretical results are validated by comparison with experimental results. The measured isolation of the 4 = 4 array is 26.5 dB at 12.0 GHz, which is obviously better than that of the classical ones. The dual-polarized array presented in this paper can be improved further by means of corrections in the feeder width and length, corporate feed design, etc. They are promising for practical applications.

ABSTRACT: We implement the Du Fort᎐Frankel modified explicit finite-difference beam propagation method (MEFD) to model threedimensional optical de¨ ices using parallel computers. Accuracy comparisons with other parallel FD᎐BPMs are made, and we obser¨ e that the MEFD is ¨ ery accurate and efficient. The parallel implementation of MEFD shows a large run-time computer sa¨ ings compared to other parallel FD᎐BPM algorithms. 䊚 2000 John Wiley & Sons, Inc. Microwave Opt Technol Lett 24: 179᎐182, 2000. Key words: beam propagation method; finite-difference analysis; modeling; numerical analysis; optical wa¨ eguide theory; partial differential equations; parallel processing implementations I. INTRODUCTION

As we come to the stage of integrating many optical processing elements onto one substrate, the optical circuit is becomContract grant sponsor: British Council

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