ators. These behaviors agree well with the parametric analysis discussed in the previous paragraphs. 3. RESULTS
The designed antenna was fabricated and experimentally studied. The performance of the antenna was measured with an 8719ES vector network analyzer and a far-field measurement system. Figure 9 compares the measured and simulated return loss characteristics of the proposed antenna. A good agreement is observed between the simulated and the measured results. Four resonant frequencies were obtained at 0.93 GHz, 1.88 GHz, 3.43 GHz, and 5.2 GHz with bandwidths of 13.9%, 11.7%, 13.1%, and 6.1%, respectively. The bandwidths adequately cover the four desired frequency bands. The measured radiation patterns are shown in Figure 10 at 0.93, 1.88, 3.43, and 5.2 GHz. The radiation patterns in the y-z plane were nearly omni-directional at the four frequencies, and those in the x-z plane were monopole-like. In some angular regions, the crosspolarization component was stronger than the copolarization field. However, as the crosspolarization component compensated for the copolarization field, the total field became uniform [14]. The measured peak gains for the antenna were ⫺0.87 dBi, ⫺0.52 dBi, 0.21 dBi, and ⫺0.75 dBi in the GSM900, DCS1900, WiMAX, and WLAN bands, respectively. The antenna gains were lower than that of stand-alone slot antennas [15, 16] because of the Ohmic losses of the termination resistance used for Radiator 2. 4. CONCLUSION
5. J.-I. Moon and S.-O. Park, Small chip antenna for 2.4/5.8-GHz Dual ISM-Band applications, IEEE Antennas Wireless Propag Lett 2 (2003), 313–315. 6. K.C. Hwang, A modified Sierpinski fractal antenna for multiband application, IEEE Antennas Wireless Propag Lett 6 (2007), 357–360. 7. W.-G. Jang and J.-H Choi, Design of a wide and multiband aperturestacked patch antenna with reflector, Microwave Opt Technol Lett 49 (2007), 2822–2824. 8. S.C. Kim, S.H. Lee, and Y.-S. Kim, Multi-band monopole antenna using meander structure for handheld terminals, Electron Lett 44 (2008), 331–332. 9. J.-S. Lee, H. Rhyu, and B. Lee, Design concept of multi-band antenna with resonant circuit on PCB, Electron Lett 43 (2007), 5– 6. 10. G.K.H. Lui and R.D. Murch, Compact dual-frequency PIFA designs using LC resonators, IEEE Trans Antennas Propag 49 (2001), 1016 – 1019. 11. W.-C. Liu, Design of a multiband CPW-fed monopole antenna using a particle swarm optimization approach, IEEE Trans Antennas Propag 53 (2005), 3273–3279. 12. J.-S. Chen, Studies of CPW-fed equilateral triangular-ring slot antennas and triangular-ring slot coupled patch antennas, IEEE Trans Antennas Propag 53 (2005), 2208 –2211. 13. Ansoft Corp. Ansoft High Frequency Structure Simulator (HFSS), Ver. 11.0, Ansoft Corp., Pittsburgh, PA. 14. D.-U. Sim and S.-O. Park, A triple-band internal antenna: Design and performance in presence of the handset case, battery, and human head, IEEE Trans Electromagn Compat 47 (2005), 658 – 666. 15. J.-Y. Jan and C.-Y. Hsiang, Wideband CPW-fed slot antenna for DCS, PCS, 3G and Bluetooth bands, Electron Lett 42 (2006), 1377–1378. 16. A.A. Omar, M.C. Scardelletti, Z.M. Hejazi, and N. Dib, Design and measurement of self-matched dual-frequency coplanar waveguide-fedslot antennas, IEEE Trans Antennas Propag 55 (2007), 223–226.
In this article, we proposed a multiband slot antenna. The proposed antenna was designed to cover the GSM900, DCS 1900, WiMAX, and 5.2 GHz WLAN frequency bands simultaneously. The antenna has a simple structure and is easy to fabricate on a substrate. To achieve multiband operation, two parasitic radiators are used. One has the same shape as the main radiator and is terminated with a 50 ⍀ load for operating in the GSM900, DCS 1900, and WiMAX bands, and another parasitic radiator with a circular shape is placed between the two hook-shaped radiators to create an additional resonance at the 5.2 GHz WLAN band. The results show that by controlling the electromagnetic coupling among the three radiators, quadruple-band operation can be achieved without increasing the total dimension of an antenna. The surface current distribution at each resonant frequency and the parametric analysis for various design parameters were investigated.
A COMPACT RECTANGULAR DIELECTRIC RESONATOR ANTENNA FOR UWB WIRELESS COMMUNICATION SYSTEMS
ACKNOWLEDGMENTS
Received 28 January 2009
This research was supported by the MKE (Ministry of Knowledge Economy), Korea, under the ITRC (Information Technology Research Center) support program supervised by the IITA (Institute of Information Technology Assessment) (IITA-2008-C1090-08010019). REFERENCES 1. X. Jing, Z. Du, and K. Gong, A compact multiband planar antenna for mobile handsets, IEEE Antennas Wireless Propag Lett 5 (2006), 343–345. 2. Z.N. Chen and M.Y.W. Chia, Broadband planar inverted-L antennas, IET Proc Microwaves Antennas Propag 148 (2001), 339 –342. 3. R.A. Bhatti and S.O. Park, Hepta-band internal antenna for personal communication handsets, IEEE Trans Antennas Propag 55 (2007), 3398 –3403. 4. G. Augustin, P.C. Bybi, V.P. Sarin, P. Mohanan, C.K. Aanandan, and K. Vasudevan, A compact dual-band planar antenna for DCS-1900/ PCS/PHS, WCDMA/IMT-2000, and WLAN applications, IEEE Antennas Wireless Propag Lett 7 (2008), 108 –111.
DOI 10.1002/mop
© 2009 Wiley Periodicals, Inc.
Mohssin Aoutoul, Otman El-Mrabet, Mohamed Essaaidi, and Ahmed El Moussaoui Faculty of Science, Electronics and Microwaves Group, Abdelmalek Essaadi University, Tetuan 93000, Morocco; Corresponding author:
[email protected]
ABSTRACT: A compact rectangular dielectric resonator antenna (DRA) is presented for ultra wide band communication systems. Ansoft HFSS 3D electromagnetic solver is used for the design optimization and measurement results are also provided. The proposed antenna has a low profile (⬃3), a small size, and a low permittivity constant (10.2). An impedance bandwidth about 46% (from 6.9 to 11 GHz) was achieved making this antenna suitable for UWB applications. © 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett 51: 2281–2286, 2009; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.24623 Key words: ultra wideband (UWB); rectangular dielectric resonator antenna (DRA); ansoft HFSS 1. INTRODUCTION
Dielectric resonator antennas (DRA) have received much attention recently owing to many attractive features, such as high radiation efficiency, considerable bandwidth, light weight, small size, and
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low profile. However, the relative bandwidth of a single DRA, typically below 10%, can not meet the increasing demand for wideband applications [i.e., Ultra wide band (UWB)]. To enhance the bandwidth of DRAs, many techniques have been proposed, such as stacking multiple dielectric resonators (DRs) [1–3], using parasitic DR elements [4], and utilizing special DR geometries [5]. However, these techniques will increase the antenna size and cost. In this article, a compact ultra wide band rectangular DRA is proposed using a simple DRA topology. This antenna is fed by a stepped microstrip line. Although the main antenna structure is similar to that reported in [6], which is less compact, the fundamental antenna working principle, as detailed in section II, is different and allows to have much wider bandwidth compared with the original design at the similar frequency band.
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This work aims to optimize the antenna given in [6] to have UWB performances and compact size. This has been achieved through a rectangular dielectric antenna printed on the widely used Rogers RT/Duroid 3010 with dimensions 30 ⫻ 45 ⫻ 1.27 mm3 as shown in Figure 1. The dimensions of this DRA (r ⫽ 10.2) are fixed at 10 ⫻ 10 ⫻ 2.5 mm3 and the dimensions of truncated ground plane are 30 ⫻ 25 mm2. The dimensions of the microstrip feed line are W1 ⫽ 1.2 mm and 11 ⫽ 19 mm whereas the width and length of the wide stripline are, respectively, W2 ⫽ 3 mm and 12 ⫽ 11 mm. There are three slots in the truncated ground plane beneath the stepped feed line with the following dimensions, respectively, WS1 ⫽ 1.2 mm, LS1 ⫽ 6 mm, WS2 ⫽ 3 mm, LS2 ⫽ 2 mm, WS3 ⫽ 4 mm, and LS3 ⫽ 2 mm. A metallic layer is placed in the
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Figure 2 The effect of WS3 before using the rectangular metallic layer
middle of the DRA button with dimensions 3.9375 ⫻ 3.9375 ⫻ 1.2 mm3 as shown in Figure 1. Therefore, the proposed antenna has a UWB performance (6.9 –11 GHz, with a fractional bandwidth of 46%) and a compact size, which allows it to be integrated in different UWB electronic systems, whereas the antenna reported in [6] is a broadband antenna (8 –9 GHz) and has a size of 90 ⫻ 90 mm2. Furthermore, the key difference between these two antennas is the fact that the ground plane has been removed beneath the proposed DRA, the fact that contributes considerably to its UWB performance. 3. PARAMETRIC STUDIES
Every geometrical parameter has different effects on the overall performances of the proposed antenna. In the following section, the effects of three parameters of the proposed antenna shown in Figure 1, namely, (i) the third slot width WS3; (ii) the DRA height; and (iii) the metallic layer height will be investigated in depth using Ansoft HFSS simulator [7]. 3.1. Third Slot Width Effect To explore the effects of the third slot width; the DRA antenna is considered without the metallic layer. Figure 2 gives its return loss
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Figure 1 The structure of the DRA under consideration: (a) 3D view, (b) bottom view illustrating the slots on the truncated ground plane, (c) side view, and (d) top view
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Figure 3 The effect of the metallic layer height
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as a function of frequency for different third slot width values. From this study, we find that the relative bandwidth is 31% spanning the frequency band from 6.86 to 9.41 GHz when WS3 ⫽ 4 mm. 3.2. Metallic Layer Effect To increase the bandwidth; a rectangular metallic layer is placed in the middle of the DRA. After several optimization tests, we found that the metallic layer length and width (LRM, WRM) have to satisfy the following relationship WRM ⫽ LRM ⫽ W2 (1 ⫹ ⌬) to get a larger bandwidth. The optimum value obtained for ⌬ is 0.3125. We have also investigated this layer’s height effect (HRM). In Figure 3, the return loss of the antenna is plotted considering a rectangular metallic layer with WRM ⫽ LRM ⫽ 3.9375 mm and for different values of its height (HRM). This figure shows that the studied DRAs lowest operating frequency remains unchanged; whereas the highest operating frequency experiences an insignificant variation. 3.3. DRA Height (HDRA) Effect In Figure 4, the return loss of the antenna is depicted as a function of frequency with the DRA height HDRA as a parameter. Following several optimization simulations, we found that HDRA must be equal to 2.0833*HRM to enhance the bandwidth of this antenna.
Figure 6 Antenna structure: (a) top view, (b) bottom view, and (c) DRA with metallic layer. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com]
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Figure 7 Return loss of the fabricated antenna with HDRA ⫽ 2.5 mm
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Figure 8
Radiation pattern at 7.23, 8.19, 9.17, and 10.44 GHz, H-Plane
Figure 9 Radiation pattern at 7.23, 8.19, 9.17, and 10.44 GHz at E-Plane, Phi ⫽ 0
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Figure 10
Radiation pattern at 7.23, 8.19, 9.17, and 10.44 GHz at E-Plane, Phi ⫽ 90
From this figure, we can see that the bandwidth upper frequency increases when HDRA increases. 3.4. Optimum Structure According to the above analyses; it is obvious that the rectangular metallic layer is the most important parameter allowing to increase the bandwidth of the proposed antenna (Figs. 2 and 3). The optimum DRA parameters values obtained by simulations are: HDRA ⫽ 3.3 mm, WRM ⫽ LRM ⫽ 3.9375 mm, HDRA ⫽ 2.0833*HRM, WS3 ⫽ 5 mm. The dielectric constant value (r ⫽ 10.2 for both DRA and substrate) is the same as the one used in literature [6]. In Figure 5, the return loss with optimum antenna parameters is depicted versus frequency using Ansoft HFSS. The HFSS simulations’ results show that the relative bandwidth achieved with optimum parameters is about 56% ranging from 6.78 GHz to 12.1 GHz. 3.5. Measurement A prototype of this antenna was fabricated with HDRA ⫽ 2.5 mm (refer Fig. 6). Because of some fabrication constraints, we were obliged to use HDRA ⫽ 2.5 mm. However, this value did not affect so much the bandwidth of the proposed UWB DRA (refer Figs. 5 and 7). Measurement results (refer Fig. 7) confirm those provided by HFSS simulations and prove that this is an UWB antenna operating between 6.9 and 11 GHz. Furthermore, the agreement between these results is excellent for the lower frequency and it is quite good for the upper frequency which shows a discrepancy less than 0.5 GHz, while the discrepancy in the behavior of the measured and simulated return loss curves between these two frequencies does not affect the bandwidth defined for the values of S11 below ⫺10 dB. Figures 8–10 illustrate the simulated and measured co-polar-
DOI 10.1002/mop
ization and cross-polarization radiation patterns of the proposed antenna in three principal planes at 7.23 GHz; 8.19 GHz; 9.17 GHz; and 10.44 GHz. In general, the simulated results are in good agreement with measurement. The radiation pattern is seen to be unstable over the operating band due to the presence of the metallic layer inside the DRA, which results in high surface current density. Figure 11 shows the antenna’s measured gain versus frequency for different values of frequencies along the operating frequency band. The radiated modes of the proposed compact UWB DR antenna are similar to those of a conventional rectangular DR antenna, given the fact that the metallic layer we have embedded in the DR is small and perpendicular to the electric field.
Figure 11 UWB DRA gain versus frequency
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4. CONCLUSION
1. INTRODUCTION
In this article, a compact UWB dielectric resonator antenna has been developed for ultra wideband communications. The introduction of the rectangular metallic layer in this antenna has allowed us to achieve a 46% bandwidth (from 6.9 to 11 GHz). The proposed antenna has a simple configuration and it is easy to fabricate. It also satisfies the ⫺10 dB return loss requirement from 6.9 to 11 GHz which makes it suitable for DS-UWB high band applications (i.e., from 6 to 10.1 GHz).
Because of the many challenges such as patient well-being, implantable devices are gaining popularity in medical applications [1,2]. Two particular domains of interest are temperature and blood pressure measurements inside a human body [3]. Most of the proposed implanted devices contain a sensor associated to transceiver unit often powered by small batteries. However, such an operating mode is not particularly suitable for a long-life implantation because the sensor is inaccessible and its life-time is limited. Some years ago, the use of passive SAW components for remote sensing was demonstrated [4], especially for temperature and pressure measurements but also for their possibility to be wirelessly interrogated [5,6] and for their totally passive technology. For example, a system to determine the body temperature of a dog has been presented in [7]. A fairly good implanted antenna is important for measurement accuracy and the total budget link which relies on the matching between the antenna and the sensor. Because of communication through the human body, the electromagnetic waves experience multiple reflections and are therefore strongly attenuated. In addition, the gain of the antenna is always severely degraded. Consequently, there is a growing interest in miniaturization of implanted antennas in dissipative media and optimization of their performance. Most of the proposed antennas have been studied for the Medical Implant Communications Service (MICS) in the 402– 405 MHz frequency band [8 –11]. Recently, Warty et al. [12] demonstrated the possibility to monitor the intracranial pressure in a skull with an implanted 4 ⫻ 5 mm2 Planar Inverted-F Antenna (PIFA) operating at 2.45 GHz. Yu et al. presented in [13] a 10-mm diameter microstrip antenna based on the short pin technique at 2.45 GHz. In Ref. 14, Karacolak et al. optimized a 22.5 mm by 22.5 mm dual-band spiral antenna for the 400-MHz MICS and the 2.45-GHz Industrial Scientific Medical (ISM) frequency bands. All the proposed structures are interesting in terms of gain or their dual-band behavior. However, their sizes are not necessarily suitable for a realistic implantation in the human body and their radiation patterns are not always omnidirectional. In this article, the proposed antenna is a 2.5 mm by 4 mm spiral antenna optimized for the 2.45-GHz ISM frequency band. This antenna will be implanted inside a human body close to the heart. The choice of such antenna over other radiating structures is also conditioned to its ability to radiate quasi-omnidirectionally due to the fact that any prior knowledge of the orientation of the interrogating unit regarding the patient is known.
ACKNOWLEDGMENT
The authors of this article thank Prof. Ahmed Kishk from the University of Mississippi and Prof. John Volakis from Ohio State University for their invaluable help and support for the fabrication and measurement of this antenna. REFERENCES 1. A.A. Kishk, B. Ahn, and D. Kajfez, Broadband stacked dielectric resonator antennas, Electron Lett 25 (1989), 1232–1233. 2. S.M. Shum and K.M. Luk, Stacked annular-ring dielectric resonator antenna excited by axi-symmeric coaxial probe, IEEE Trans Antennas Propag 43 (1995), 889 – 892. 3. A. Petosa, N. Simons, R. Siushansian, A. Ittipiboon, and M. Cuhaci, Design and analysis of multisegment dielectric resonator antennas, IEEE Trans Antennas Propag 48 (2000), 738 –742. 4. R.N. Simons and R.Q. Lee, Effect of parasitic dielectric resonators on CPW/aperture-coupled dielectric resonator antennas, Proc Inst Elect Eng 140 (1993), 336 –338. 5. A.A. Kishk, Y. Yan, and A.W. Gilson, Conical dielectric resonator antennas for wide-band applications, IEEE Trans Antennas Propag 50 (2002), 469 – 474. 6. M. Saed and R. Yadla, Microstrip-fed low profile and compact dielectric resonator antenna, Prog Electromag Res PIER 56 (2006), 151–162. 7. Ansoft Corp., Pittsburgh, PA 15219. © 2009 Wiley Periodicals, Inc.
SMALL ELECTRICAL ANTENNA FOR SAW SENSOR BIOTELEMETRY Gwladys Collin, Ali Chami, Cyril Luxey, Philippe Le Thuc, and Robert Staraj Laboratoire d’Electronique, Antennes et Te´le´communications, Universite´ de Nice-Sophia Antipolis/UMR-CNRS, Baˆt.4, 250, rue Albert Einstein, 06560 Valbonne, France; Corresponding author:
[email protected]
2. WIRELESS SYSTEM AND IMPLANTED STRUCTURE Received 28 January 2009 ABSTRACT: A spiral antenna matched to a SAW sensor for blood pressure and body temperature monitoring is proposed. As the antenna/ SAW device is dedicated to be implanted in a human body, a small antenna with omnidirectional pattern is required. The proposed structure has a total area of 4 ⫻ 2.5 mm2 and a 1.1 mm height. To take into account the human body effect, we use a homogenous phantom with a dielectric permittivity of 52 and a conductivity of 1.74 S/m. The ⫺10 dB impedance bandwidth of the proposed antenna is 66% from 2.22 to 3.83 GHz; moreover, a quasi-isotropic radiation pattern is achieved. © 2009 Wiley Periodicals, Inc. Microwave Opt Technol Lett 51: 2286 –2293, 2009; Published online in Wiley InterScience (www.interscience.wiley. com). DOI 10.1002/mop.24622 Key words: spiral antenna; phantom; passive implanted medical device; SAW sensor
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The wireless system is composed of an interrogator unit located outside the human body (represented by a square box in Fig. 1) and an implanted device made up of the association of two SAW sensors and two antennas. A SAW sensor is composed of an InterDigital Transducer (IDT) and some reflectors placed onto a piezoelectric substrate. Such devices use the piezoelectric effect to convert an incoming electromagnetic wave into a mechanical wave. Changes in temperature or pressure modify the elastic constant of the piezoelectric substrates which induces a modification of the velocity of the mechanical wave propagation. There are two different types of SAW sensors that allow the determination of different physical quantities: resonator sensors and delay line sensors. SAW resonator technology is used in this work. A resonator can be modeled by a Butterworth–Van Dyke equivalent circuit. In this model, the circuit is composed of two
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DOI 10.1002/mop
